Frequency selective transmit signal weighting for multiple antenna communication systems

ABSTRACT

A method for communication includes performing by one or more processors and/or circuits in a communication device functions including determining transmit power weights, which are to be utilized for communicating one or more wireless signals via a wireless channel, as a function of frequency of a wireless signal communicated via the wireless channel. The determining may be based on a transmission mode of the wireless signal and a state of the wireless communication channel. Transmit antenna spatial weights may be determined for communicating the one or more wireless signals via a plurality of antennas. The one or more wireless signals may be weighted with the transmit power weights and/or one of the transmit antenna spatial weights. The weighted one or more wireless signals may be transmitted via one of a plurality of antennas in accordance with the transmission mode.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.12/468,722 (now U.S. Pat. No. 7,869,537) filed May 19, 2009, which is acontinuation of U.S. patent application Ser. No. 10/903,876 (now U.S.Pat. No. 7,535,969) filed Jul. 29, 2004, which claims priority under 35U.S.C. §119(e) to U.S. provisional application Ser. No. 60/491,128,entitled FREQUENCY SELECTIVE TRANSMIT SIGNAL WEIGHTING FOR MULTIPLEANTENNA SYSTEMS, filed Jul. 29, 2003, which is herein incorporated byreference in its entirety. This application is also related to U.S.non-provisional application Ser. No. 10/801,930 (now U.S. Pat. No.7,822,140), entitled MULTI-ANTENNA COMMUNICATION SYSTEMS UTILIZINGRF-BASED AND BASEBAND SIGNAL WEIGHTING AND COMBINING, filed Mar. 16,2004, to U.S. non-provisional application Ser. No. 10/835,255 (now U.S.Pat. No. 7,539,274), entitled WEIGHT GENERATION METHOD FOR RF SIGNALCOMBINING IN MULTI-ANTENNA COMMUNICATION SYSTEMS, filed Apr. 29, 2004,and to copending U.S. non-provisional application Ser. No. 10/886,510,entitled SYSTEM AND METHOD FOR RF SIGNAL COMBINING AND ADAPTIVE BITLOADING FOR DATA RATE MAXIMIZATION IN MULTI-ANTENNA COMMUNICATIONSYSTEMS, filed Jul. 7, 2004, which itself claims priority to U.S.provisional Ser. No. 60/485,915 filed Jul. 9, 2003, all of which areincorporated by reference and assigned to the assignee of the presentapplication.

FIELD OF THE INVENTION

The present invention relates to communication systems utilizingtransmitters and receivers having multiple antenna elements. Moreparticularly, the present invention relates to a weight generationmethod for facilitating signal weighting and combining, withinmulti-antenna systems.

BACKGROUND OF THE INVENTION

Most current wireless communication systems are composed of nodesconfigured with a single transmit and receive antenna. However, for awide range of wireless communication systems, it has been predicted thatthe performance, including capacity, may be substantially improvedthrough the use of multiple transmit and/or multiple receive antennas.Such configurations form the basis of many so-called “smart” antennatechniques. Such techniques, coupled with space-time signal processing,can be utilized both to combat the deleterious effects of multipathfading of a desired incoming signal and to suppress interfering signals.In this way both the performance and capacity of digital wirelesssystems in existence or being deployed (e.g., CDMA-based systems,TDMA-based systems, WLAN systems, and OFDM-based systems such as IEEE802.11a/g) may be improved.

The impairments to the performance of wireless systems of the typedescribed above may be at least partially ameliorated by usingmulti-element antenna systems designed to introduce a diversity gain andsuppress interference within the signal reception process. This has beendescribed, for example, in “The Impact of Antenna Diversity On theCapacity of Wireless Communication Systems”, by J. H. Winters et al,IEEE Transactions on Communications, vol. 42, No. 2/3/4, pages1740-1751, February 1994. Such diversity gains improve systemperformance by mitigating multipath for more uniform coverage,increasing received signal-to-noise ratio (SNR) for greater range orreduced required transmit power, and providing more robustness againstinterference or permitting greater frequency reuse for higher capacity.

Within communication systems incorporating multi-antenna receivers, itis known that a set of M receive antennas are capable of nulling up toM-1 interferers. Accordingly, N signals may be simultaneouslytransmitted in the same bandwidth using N transmit antennas, with thetransmitted signal then being separated into N respective signals by wayof a set of N antennas deployed at the receiver. Systems of this typeare generally referred to as multiple-input-multiple-output (MIMO)systems, and have been studied extensively. See, for example, “Optimumcombining for indoor radio systems with multiple users,” by J. H.Winters, IEEE Transactions on Communications, Vol. COM-35, No. 11,November 1987; “Capacity of Multi-Antenna Array Systems In IndoorWireless Environment” by C. Chuah et al, Proceedings of Globecom '98Sydney, Australia, IEEE 1998, pages 1894-1899 November 1998; and “FadingCorrelation and Its Effect on the Capacity of Multi-Element AntennaSystems” by D. Shiu et al, IEEE Transactions on Communications vol. 48,No. 3, pages 502-513 March 2000.

One aspect of the attractiveness of multi-element antenna arrangements,particularly MIMOs, resides in the significant system capacityenhancements that can be achieved using these configurations. Under theassumption of perfect estimates of the applicable channel at thereceiver, in a MIMO system with N transmit and N receive antennaelements, the received signal decomposes to N “spatially-multiplexed”independent channels. This results in an N-fold capacity increaserelative to single-antenna systems. For a fixed overall transmittedpower, the capacity offered by MIMOs scales linearly with the number ofantenna elements. Specifically, it has been shown that with N transmitand N receive antennas an N-fold increase in the data rate over a singleantenna system can be achieved without any increase in the totalbandwidth or total transmit power. See, e.g., “On Limits of WirelessCommunications in a Fading Environment When Using Multiple Antennas”, byG. J. Foschini et al, Wireless Personal Communications, Kluwer AcademicPublishers, vol. 6, No. 3, pages 311-335, March 1998. In experimentalMIMO systems predicated upon N-fold spatial multiplexing, more than Nantennas are often deployed at a given transmitter or receiver. This isbecause each additional antenna adds to the diversity gain and antennagain and interference suppression applicable to all Nspatially-multiplexed signals. See, e.g., “Simplified processing forhigh spectral efficiency wireless communication employing multi-elementarrays”, by G. J. Foschini, et al, IEEE Journal on Selected Areas inCommunications, Volume: 17 Issue: 11, November 1999, pages 1841-1852.

Although increasing the number of transmit and/or receive antennasenhances various aspects of the performance of MIMO systems, thenecessity of providing a separate RF chain for each transmit and receiveantenna increases costs. Each RF chain is generally comprised of a lownoise amplifier, filter, downconverter, and analog to digital toconverter (A/D), with the latter three devices typically beingresponsible for most of the cost of the RF chain. In certain existingsingle-antenna wireless receivers, the single required RF chain mayaccount for in excess of 30% of the receiver's total cost. It is thusapparent that as the number of transmit and receive antennas increases,overall system cost and power consumption may dramatically increase. Itwould therefore be desirable to provide a technique for utilizingrelatively larger numbers of transmit/receive antennas withoutproportionately increasing system costs and power consumption.

The above-referenced copending non-provisional application Ser. No.10/801,930 provides such a technique by describing a wirelesscommunication system in which it is possible to use a smaller number ofRF chains within a transmitter and/or receiver than the number oftransmit/receiver antennas utilized.

In the case of an exemplary receiver implementation, the signal providedby each of M (M>N) antennas is passed through a low noise amplifier andthen split, weighted and combined in the RF domain with the signals fromthe other antennas of the receiver. This forms NRF output signals, whichare then passed through NRF chains. The output signals produced by anA/D converter of each RF chain are then digitally processed to generatethe N spatially-multiplexed output signals. By performing the requisiteweighting and combining at RF using relatively inexpensive components,an N-fold spatially-multiplexed system having more than N receiveantennas, but only NRF chains, can be realized at a cost similar to thatof a system having N receive antennas. That is, receiver performance maybe improved through use of additional antennas at relatively low cost. Asimilar technique can be used within exemplary transmitterimplementations incorporating NRF chains and more than N transmitantennas.

The RF-based weighting techniques described in the above-referencedcopending non-provisional application Ser. No. 10/801,930 advantageouslyenable the same type of combining of spatially weighted signals to beperformed in the RF domain as is done at baseband. One advantage ofthese techniques is that RF weighting and combining may be performedusing only N transmit and N receive RF chains, independent of the numberof transmit and receive antennas. Furthermore, notwithstanding the factthat the '930 application describes RF-based weighting and combining, itremains possible to implement the digital signal processing schemesprior to conversion to analog/RF within the transmitter and subsequentto conversion to digital from analog/RF within the receiver. Suchtechniques may include successive interference cancellation in the caseof MIMO systems (see, e.g., “V-BLAST: An architecture for realizing veryhigh data rates over the rich-scattering wireless channel,” inProceedings of URSI ISSSE, September, 1998, pp. 295-300).

Although the techniques described in the above-referenced copendingnon-provisional application Ser. No. 10/801,930 may not offerperformance identical to baseband techniques in the case of temporaland/or frequency domain signal processing, it may still be preferable toemploy such techniques as a result of the lower costs involved.Frequency domain processing is used in systems in which, for example,the transmitted signal consists of a number of frequency subcarriers.This type of signal processing is required to be performed whenimplementing systems based upon orthogonal frequency divisionmultiplexing (OFDM), such as the wireless local area network systemspopularly referred to simply as “802.11(a)” and “802.11(g)”.Alternatively, for the same or lower cost as is required by conventionalapproaches, the techniques of the above-referenced copendingnon-provisional application Ser. No. 10/801,930 may be employed toenable the use of a greater number of antennas, which may result insubstantially superior performance relative to such conventionalapproaches.

SUMMARY OF THE INVENTION

The present invention is directed to a system and method for generatingtransmit signal weight values with frequency for weighting elementsincluded within the signal weighting and combining arrangements used invarious multi-antenna transmitter and receiver structures. Specifically,the present invention may be used in conjunction with RF-based weightingand combining arrangements within multi-antenna transmitter and receiverstructures disposed to process one or more information signals modulatedupon respective pluralities of subcarrier signals. The present inventionmay also find application when baseband weighting and combiningarrangements are incorporated within the same multi-antenna transmitteror receiver structure, or furthermore when both RF-based and basebandweighting and combining arrangements are used.

Consistent with the invention, the frequency-selective weight generationmethod varies with the transmission mode. The inventive weightgeneration method may be employed within several different types ofmulti-antenna communication systems including, for example, thosedescribed within the above-referenced copending non-provisionalapplications. In particular embodiments the inventive technique may beapplied to a multi-antenna receiver within a “single channel” (SC)system (i.e., a system lacking spatial multiplexing), to a multi-antennatransmitter in a single channel system, or to a MIMO system employingspatial multiplexing.

As is described herein, the frequency-selective transmit signal weightvalues can be generated based on the transmit and receive spatialweights to optimize a performance measure such as the outputsignal-to-noise ratio, the output bit error rate, or the output packeterror rate of the multi-antenna communication system. Thefrequency-selective transmit signal weight values can also be generatedjointly with transmit and receive spatial weights to optimizeperformance.

In one aspect, the present invention relates to a method, and means toimplement the method, for transmitting a signal over a wireless channel.The method including the steps of: acquiring information representativeof a state of the channel based upon an initial signal transmitted overthe channel; acquiring information representative of a transmission modeof the signal; determining transmit weighting values for the signal as afunction of frequency based upon the state of the channel and thetransmission mode of the signal; weighting the signal with the transmitweighting values thereby generating a weighted signal; and transmittingthe weighted signal in accordance with the transmission mode.

The invention is also directed to a system for transmitting a signal.The system includes a receiver configured to receive an initial signaland estimate a channel state information for the initial signal; and atransmitter configured to transmit the signal according to atransmission mode and to perform transmit signal weighting as a functionof frequency based on the channel state information and the transmissionmode.

In another aspect, the invention pertains to a system that includes areceiver configured to receive an initial signal and estimate acombination of transmit signal weighting values as a function offrequency based on a pre-determined channel state information andtransmission mode. The system also includes a transmitter configured toweight the signal with the said transmit weighting values therebygenerating a weighted signal and transmit the weighted signal inaccordance with the transmission mode.

BRIEF DESCRIPTION OF THE DRAWINGS

For a better understanding of the nature of the features of theinvention, reference should be made to the following detaileddescription taken in conjunction with the accompanying drawings, inwhich:

FIG. 1 is a block diagram illustratively representing a conventionalMIMO communication system;

FIG. 2 shows a block diagram of a MIMO communication system having atransmitter and a receiver configured to effect RF-based weighting andcombining;

FIG. 3 depicts the transmitter and receiver structure in asingle-channel (SC) single-input-multiple-output (SIMO) single carriersystem with a baseband combining arrangement;

FIG. 4 depicts the transmitter and receiver structure in a SC-SIMOsingle carrier system utilizing an exemplary embodiment of the inventivetransmit signal weight generation method together with a basebandcombining arrangement;

FIG. 5A depicts the transmitter and receiver structure of a SC-MIMO-OFDMsystem utilizing an embodiment of the inventive transmit signal weightgeneration method together with a baseband combining arrangement;

FIG. 5B depicts the transmitter and receiver structure of a SC-MIMO-OFDMsystem utilizing the inventive transmit signal weight generation methodtogether with an RF-based weighting and combining arrangement;

FIG. 6 is a flowchart depicting steps carried out in accordance with oneembodiment of the inventive transmit signal weight generation method;

FIG. 7 illustratively represents comparative performance packet errorrate (PER), as a function of signal-to-noise ratio (SNR) for a codedoperative mode of a SC-MIMO-OFDM system.

FIG. 8 illustratively represents comparative performance PER, as afunction of SNR for an uncoded operative mode of a SC-MIMO-OFDM system.

FIG. 9 illustratively represents comparative performance PER, as afunction of SNR for a coded 16QAM-modulated operative mode of aSC-MIMO-OFDM system.

DETAILED DESCRIPTION OF THE INVENTION

As is discussed below, the present invention according to severalembodiments is directed to a method of frequency selective,transmit-signal weight-values generation that is applicable to a varietyof communication systems that employ a variety of coding and modulationtechniques. In accordance with the invention, the weight values can begenerated to improve one or more of the communication system'sperformance characteristics including output signal-to-noise ratio andpacket error rate. Advantageously, in some embodiments the frequencyselective transmit signal weighting techniques described herein areemployed in connection with other performance enhancing techniquesincluding, but not limited to, spatial weighting, which is used toimprove performance in multi-antenna systems as described in theabove-referenced copending non-provisional application Ser. No.10/801,930, incorporated herein by reference.

In order to facilitate appreciation of the principles of the invention,an overview is provided that generally covers variousperformance-enhancing schemes utilized in connection with the frequencyselective transmit signal weighting method in accordance with severalembodiments of the present invention. This overview is followed by adetailed description of the inventive method of transmit signal weightgeneration, which may be applied, without limitation, within the contextof these and other performance enhancing schemes as disclosed herein.

I. Performance Enhancing Methodologies Spatial Multiplexing

As is known, spatial multiplexing (SM) provides a mode of signaltransmission predicated upon the use of multiple antennas at both atransmitter and a receiver in such a way that the bit rate of a wirelessradio link may be increased without correspondingly increasing power orbandwidth consumption. In the case in which N antennas are used at botha transmitter and a receiver, an input stream of information symbolsprovided to the transmitter is divided into N independent substreams.Spatial multiplexing contemplates that each of these substreams willoccupy the same “channel” (e.g., time slot, frequency, or code/keysequence) of the applicable multiple-access protocol. Within thetransmitter, each substream is separately applied to the N transmitantennas and propagated over an intervening multipath communicationchannel to a receiver. The composite multipath signals are then receivedby a receive array of N receive antennas deployed at the receiver. Atthe receiver, a “spatial signature” defined by the N phases and Namplitudes arising at the receive antenna array for a given substream isthen estimated. Signal processing techniques are then applied in orderto separate the received signals, which permits the original substreamsto be recovered and synthesized into the original input symbol stream.The principles of spatially-multiplexed communication and exemplarysystem implementations are further described in, for example, “Optimumcombining for indoor radio systems with multiple users”, by J. H.Winters, IEEE Transactions on Communications, Vol. COM-35, No. 11,November 1987, which is incorporated herein by reference in itsentirety.

Conventional MIMO System

The principles of the present invention may be more fully elucidated byfirst considering a conventional MIMO communication system, which isillustratively represented by FIG. 1. As shown, the MIMO system 100 ofFIG. 1 includes a transmitter 110 depicted in FIG. 1A and a receiver 130depicted in FIG. 1B. The transmitter 110 and receiver 130 include a setof T transmit RF chains and a set of R receive RF chains, respectively,which are configured to transmit and receive a group of Nspatially-multiplexed signals. Within the system 100 it is assumed thateither (i) T is greater than N and R is equal to N, (ii) T is equal to Nand R is greater than N, or (iii) both T and R are greater than N.

Referring to FIG. 1A, an input signal S to be transmitted, whichtypically consists of a stream of digital symbols, is demultiplexed bydemultiplexer 102 into N independent substreams S_(1, 2 . . . , N). Thesubstreams S_(1, 2 . . . , N) are then sent to digital signal processor(DSP) 105, which generates a set of T output signals T_(1, 2 . . . , T).The T output signals T_(1, 2 . . . , T) are typically generated from theN substreams S_(1, 2 . . . , N) by weighting, i.e., multiplying by acomplex number, each of the N substreams S_(1, 2 . . . , N) by Tdifferent weighting coefficients to form NT substreams. These N·Tsubstreams are then combined in order to form the T output signalsT_(1, 2 . . . , T). The T output signals T_(1, 2 . . . , T) are thenconverted to T analog signals A_(1, 2 . . . , T) using a set of Tdigital-to-analog (D/A) converters 108. Each of the T analog signalsA_(1, 2 . . . , T) is then upconverted to the applicable transmitcarrier RF frequency within a mixer 112 by mixing with a signal providedby a local oscillator 114. The resulting set of T RF signals (i.e.,RF_(1, 2 . . . , T)) are then amplified by respective amplifiers 116 andtransmitted by respective antennas 118.

Referring now to FIG. 1B, the RF signals transmitted by the transmitter110 are received by a set of R receive antennas 131 deployed at thereceiver 130. Each of the R signals received by an antenna 131 isamplified by a respective low noise amplifier 133 and passed through afilter 135. The resultant filtered signals are then each downconvertedfrom RF to baseband using mixers 137, each of which is provided with asignal from local oscillator 138. Although the receiver of FIG. 1B isconfigured as a homodyne receiver, a heterodyne receiver characterizedby an intermediate IF frequency could also be used. The respective Rbaseband signals produced by the mixers 137 are then converted todigital signals using a corresponding set of R analog-to-digital (A/D)converters 140. The resulting R digital signals D_(1, 2 . . . , R) arethen weighted and combined using digital signal processor 142 to form Nspatially-multiplexed output signals S′_(1, 2 . . . , N), which compriseestimates of the transmitted signals S_(1, 2 . . . , N). The N outputsignals S′_(1, 2 . . . , N) are then multiplexed using a multiplexer 155in order to generate an estimate 160 (S′) of the original input signalS.

RF Weighting and Combining in Spatially-Multiplexed CommunicationSystems

Turning now to FIG. 2, there is shown a block diagram of a MIMOcommunication system 200 having a transmitter 210 and receiver 250configured in accordance with the principles of the above-referencednon-provisional patent applications. In the implementation of FIG. 2,the transmitter 210 and receiver 250 effect N-fold spatial multiplexingusing only N transmit/receive RF chains, even though more than Ntransmit/receive antennas are respectively deployed at the transmitter210 and receiver 250. Specifically, the transmitter 210 includes a setof MT transmit antennas 240 and the receiver includes a set of MRreceive antennas 260, it being assumed that either (i) MT is greaterthan N and MR is equal to N, (ii) MT is equal to N and MR is greaterthan N, or (iii) both MT and MR are greater than N.

As shown in FIG. 2A, an input signal S to be transmitted isdemultiplexed by demultiplexer 202 into N independent substreamsSS_(1, 2 . . . , N). The substreams SS_(1, 2 . . . , N) are thenconverted to N analog substreams AS_(1, 2 . . . , N) using acorresponding set of D/A converters 206. Next, the N analog substreamsAS_(1, 2 . . . , N) are upconverted to the applicable transmit carrierRF frequency using a set of mixers 212 provided with the signal producedby a local oscillator 214. The resultant NRF signals (i.e.,RF_(1, 2 . . . , N)) are then each split MT ways by dividers 218 inorder to form N·(MT) RF signals. These N·(MT) RF signals are eachweighted using complex multipliers 226 _(x,y), where x identifies asignal origination point at one of the N dividers 218 and y identifies acorresponding signal termination point at one of a set of MT combiners230. The weighted RF signals are combined using the combiners 230,thereby yielding a set of MT output signals. A corresponding set of MTamplifiers 234 then amplify these MT output signals, with the amplifiedoutput signals then being transmitted using the MT antennas 240. Theweighting values of the complex multipliers 226 _(x,y) may be generatedso as to maximize the SNR or to minimize the BER of the output signal atthe receiver as described in the above-referenced U.S. non-provisionalapplication Ser. No. 10/801,930, which is incorporated by reference.

Referring to FIG. 2B, the MT RF signals transmitted by the transmitter210 are received by the set of MR receive antennas 260 deployed at thereceiver 250. Each of the MR received signals is amplified by arespective low noise amplifier 264 and then split N ways by one of a setof MR dividers 268. The resulting MR·(N) split signals are then eachweighted by respective weighting circuits 272 _(x,y), where x identifiesa signal origination point at one of the MR dividers 268 and yidentifies a corresponding signal termination point at one of a set of Ncombiners 276. These weighted signals are then combined using the Ncombiners 276 in order to form a set of N signals, which are passedthrough a corresponding set of N filters 280. The resulting N filteredsignals are then downconverted to baseband using a set of N mixers 282,each of which is provided with a carrier signal produced by a localoscillator 284. Although the receiver 250 is realized as a homodynereceiver in the embodiment of FIG. 2B, it could also be implemented as aheterodyne receiver characterized by an intermediate IF frequency. The Nbaseband signals produced by the mixers 282 are then converted todigital signals via a corresponding set of N A/D converters 286. The Ndigital signals are then further processed using digital signalprocessor 288 to form the N spatially-multiplexed output signalsSS'_(1, 2 . . . , N), which are the estimates of the N independentsubstreams SS_(1, 2 . . . , N). The N output signalsSS'_(1, 2 . . . , N) are then multiplexed via a multiplexer 292 in orderto generate the output signal S′, which is an estimate of the inputsignal S.

It is observed that the transmitter 210 and receiver 250 are capable ofimplementing, within the RF domain, the same spatial weighting or linearcombining schemes as are conventionally implemented at baseband via thesystem 100 of FIG. 1. However, the DSP 288 within the receiver 250 maystill perform many other baseband signal processing operationspotentially effected within the system 100, such as, for example,successive interference cancellation (see, e.g., “V-BLAST: Anarchitecture for realizing very high data rates over the rich-scatteringwireless channel”, Proceedings of URSI ISSSE, September 1998, pp.295-300). Again, it is a feature of the disclosed system that only Ntransmit/receive RF chains need be employed, even when substantiallymore than N transmit/receive antennas are deployed.

Coding and Equalization

Referring to FIG. 3, shown is a block diagram of a single carrier system300 using a transmitter 302 with one transmit antenna and a receiver 310with two receive antennas and a maximum likelihood sequence estimation(MLSE) equalizer and decoder 316. The signal S could be, e.g., a GSMsignal. The channel represents an environment that produces frequencyselective fading, e.g., a multipath environment typical of indoor andoutdoor propagation paths.

As shown in FIG. 3, when the signal S is transmitted over the channel,the signal S is subject to frequency selective fading that potentiallydegrades the signal S. To help offset the effects of frequency-selectivefading, the receiver ideally is a spatial-temporal whitened matchedfilter, which is matched to the fading. As shown in FIG. 3 this filterconsists of a linear equalizer 312 with a length corresponding to thechannel (plus transmit filter) memory on each antenna followed bycombining 314, with the combined signal fed into the MLSE equalizer anddecoder 316.

The encoder 304 generally represents various coding schemes, e.g.,convolutional codes, linear block codes, turbo codes or trellis codes,which may be used to enhance system performance. The followingreferences, incorporated herein by reference, provide further detail, inthe context of Enhanced Data for Global Evolution (EDGE) systems, of howfor example, varying the coding rate of a convolutional code can enhancesystem performance: “System Performance of EDGE, a Proposal for EnhancedData Rates in Existing Digital Cellular Systems”, by A. Furuskar, et al,48^(th) IEEE Vehicular Technology Conference, Volume: 2, May 1998, pages1284-1289 and “Radio Interface Performance of EDGE, a Proposal forEnhanced Data Rates in Existing Digital Cellular Systems”, by A.Furuskar, et al, 48^(th) IEEE Vehicular Technology Conference, Volume:2, May 1998, pages 1064-1068.

Waterfilling

In all the above antenna systems, a received signal still suffers fromdistortion due to frequency selective fading. With respect to systemsemploying coding schemes, e.g., the system 300 described with referenceto FIG. 3, it is known that with such distortion the capacity of thesystem can be maximized by waterfilling, see e.g., “Multiuserspatio-temporal coding for wireless communications”, Wang, J.; Yao, K.,Wireless Communications and Networking Conference, 2002, 17-21 Mar.2002, pages: 276-279 vol. 1, whereby the transmit power at eachfrequency is proportional to the channel gain at that frequency, but nopower is transmitted at a given frequency if the channel gain is below agiven threshold. However, capacity is an idealized quantity that is abound that may not be achievable because it requires perfect codingand/or equalization. In practice, the equalizer is not ideal and limitedcoding (or even no coding) is used.

Smoothing

On the other hand, without coding and equalization at the receiver,e.g., if the linear equalizer is replaced by a single complex weight andan MLSE is not used, a frequency selective transmit signal weightingtechnique referred to herein as “smoothing” may be employed to removeintersymbol interference at the receiver. With the smoothing technique,the transmit signal at each frequency is weighted with the inverse ofthe channel response at that frequency, i.e., pre-equalization thatcompensates for the frequency-selective channel is carried out. Thistechnique is similar to Tomlinson precoding, see, e.g., “New AutomaticEqualiser Employing Modulo Arithmetic”, M. Tomlinson, ElectronicsLetters, Mar. 25, 1971, vol. 7, Nos. 5/6) incorporated herein byreference. It should be noted that this is the opposite of waterfillingas described above.

It should also be noted that with perfect interleaving/deinterleavingand coding, the system performance is based on the average receivesignal-to-noise ratio (SNR). Smoothing, however, reduces this averageSNR. As a consequence, smoothing is generally not desired when a systemis utilizing coding. This is in contrast to waterfilling, which, underthese circumstances, increases the SNR.

Since many systems today use a mixture of coding and equalization, andcan operate with a variety of predetermined modes that have differentcoding and modulation techniques, with the modes changing, neither ofthese frequency selective transmit signal weighting methods is optimaland a different method for weight generation is needed.

The net effect is that the transmit filtering (frequency-selectivesignal weighting) that optimizes the performance depends not only on theequalizer used, but also on the coding and modulation technique. Sincemany systems, such as Enhanced Data for Global Evolution (EDGE) systems,operate using multiple modes, with different coding and modulation ratesfor each mode, the method for frequency selective transmit signalweighting according to several embodiments of the present invention, isbased not only on the channel state information, but also on the mode.These weights can be generated to optimize a performance measure, suchas output SNR, output bit error rate, or packet error rate.

II. Frequency Selective Transmit Signal Weighting Method

In an exemplary embodiment, the present invention relates to a frequencyselective transmit signal weight-value generation method for transmitsignal weighting in a multi-antenna communication system predicated uponimproving system performance.

The teachings of the present invention are applicable to, for example,(i) receivers using multiple antennas in what are referred to herein assingle channel systems (i.e., systems lacking spatial multiplexing),(ii) transmitters using multiple antennas in single channel systems, and(iii) systems in which a smaller number of RF chains are used at thetransmitter and/or receiver than the number of transmit/receiverantennas in a MIMO system with spatial multiplexing or single-channel.

Although the present invention may be utilized in the development ofRF-based spatial weighting and combining schemes implemented usinglow-cost RF components, the teachings of the present invention areequally applicable to implementations containing both RF-based andbaseband spatial weighting and combining arrangements. Accordingly, thepresent invention is explained hereinafter both in the context ofRF-based and baseband spatial weighting and combining schemes, which mayboth be simultaneously incorporated in various embodiments of theinvention.

In another aspect, the present invention may be used for frequencyselective transmit signal weight generation in amultiple-input-multiple-output communication system using a transmitterbroadcasting a plurality (N) of spatially-multiplexed signals (using atleast N transmit antennas), where the number of received antennas (M) isgreater than the number of spatially-multiplexed signals. The receivedsignals are split, weighted and combined at RF usingfrequency-independent weights to form a number of output signals equalto the number of spatially-multiplexed signals. The output signals arethen fed to corresponding RF chains for processing at baseband.

Exemplary Scenarios

The frequency selective transmit-signal weighting and weight-valuegeneration techniques of the present invention will be describedhereinafter with reference to the exemplary scenarios illustrativelyrepresented by FIGS. 4-9. Although the inventive method is describedherein with reference to exemplary system types, it should be recognizedthat the frequency selective transmit signal weighting method is notlimited to the specific system types described with reference to FIGS.4-9. For example, the inventive method described herein applies, withoutlimitation to the following four scenarios: 1) a receiver using multipleantennas in a single channel SIMO system without spatial multiplexing,2) a transmitter using multiple antennas in a single channelmultiple-input single output (MISO) system without spatial multiplexing,3) a transmitter using multiple antennas and a receiver using multipleantennas in a single channel MIMO system without spatial multiplexing,and 4) a system whereby a smaller number of RF chains are used at thetransmitter and/or receiver than the number of transmitter/receiverantennas in a MIMO system with spatial multiplexing.

It should also be recognized that the frequency selective transmitsignal weighting method described herein applies to the four abovedescribed system types when combined with baseband combiningarrangements, RF-based weighting and combining arrangements, as well aswith both RF-based and baseband arrangements.

For illustrative purposes, many of the following examples are describedwith reference to systems utilizing OFDM modulation; however, thefrequency selective transmit signal weighting method described herein insome embodiments is applied to systems based upon a direct sequencespread spectrum (DS-SS). The above-referenced copending U.S.non-provisional application Ser. No. 10/801,930, describes in greaterdetail several such systems to which the present frequency selectivetransmit signal weighting method is applicable.

Referring first to FIG. 4, shown is a block diagram of a single carriersystem 400 using one transmit antenna 402 and a receiver 404 with tworeceive antennas 406 _(A), 406 _(B) in accordance with one embodiment ofthe present invention. As shown, the transmitter 408 in the presentembodiment includes an encoder block 410, a channel state information(CSI) and mode portion 412, a weight calculation portion 414 and asignal weighting portion 416. The receiver in the present embodimentincludes a maximum likelihood sequence estimation (MLSE) equalizer 418.

The signal S could be, e.g., a GSM signal, but this is certainly notrequired. With frequency-selective fading, the receiver 404 in someembodiments includes a spatial-temporal whitened matched filter, whichis matched to the fading. As shown in FIG. 4 this filter includes alinear equalizer 420 _(A), 420 _(B) with a length corresponding to thechannel (plus transmit and receive filters e.g., Root Raised Cosinefilter) memory on each antenna followed by combining, with the combinedsignal 424 fed into the MLSE equalizer 418 and decoder. With such areceiver 404, assuming that total transmit power is fixed and perfectchannel state information is available at the transmitter 408, thecapacity of the system 400 is maximized by waterfilling, i.e., thetransmit power at each frequency is proportional to the channel gain (orno power if the channel gain is below a threshold).

As previously discussed, however, capacity is based on ideal coding, andthus, the waterfilling solution only applies with ideal coding andequalization. In practice, the equalizer is not ideal, and limitedcoding (or even no coding) is often used.

Without coding and temporal equalization, i.e., if the linear equalizer420 _(A), 420 _(B) is replaced by a single complex weight and an MLSE418 is not used, then the performance is optimized by the smoothingapproach discussed with reference to FIG. 3.

Further, it should be noted that with perfectinterleaving/deinterleaving and coding, the system performance is basedon the average receive signal-to-noise ratio (SNR). This average SNR isreduced by smoothing and increased by waterfilling.

Thus, as discussed further herein, the weight calculation portion 414 ofthe transmitter 408 determines transmit signal weighting (also referredto as transmit filtering) as a function of not only the equalizer used,but also of the coding and modulation technique. Beneficially, theweight calculation portion 414 produces weight values based on channelstate information and mode, and it is adaptable to many systems, such asEDGE systems, that operate using multiple modes, with different codingand modulation rates for each mode. In several embodiments, theseweights are generated to optimize a performance measure, such as outputSNR or packet error rate.

Although FIG. 4 describes some embodiments of the present invention inthe context of a single carrier system, such as GSM, the frequencyselective transmit signal weighting according to several embodiments canalso be applied to CDMA or WCDMA systems, where a RAKE receiver would beused. It can also be used with an OFDM system, and results for atwo-transmit and two receive antenna single-channel MIMO system for theWLAN OFDM system 802.11a are described further herein.

In several embodiments, the frequency selective transmit signal weightvalues and transmit/receive spatial weight values are jointlycalculated. Techniques to calculate the spatial weights under a varietyof scenarios have been disclosed within the above-referenced co-pendingU.S. non-provisional application Ser. No. 10/801,930 (now U.S. Pat. No.7,539,274), with the spatial weight values calculated to improve (e.g.,optimize), such performance measures as output SNR and packet errorrate.

In several embodiments of the present invention, a global searchtechnique is utilized to identify the frequency selective transmitsignal weights that improve performance measures. This techniqueincludes searching a table for combinations of frequency selectivetransmit signal weight values for a given transmission mode that improveperformance measures including, for example, SNR and BER values. In someof these embodiments, both transmit signal weights and transmit/receivespatial weights are identified through this global search.

For example, a search engine looks for the combination of RF/baseband(i.e. spatial) weight values and transmit signal weights, which jointlyimproves (e.g., optimizes) a given criterion (e.g., max. SNR, min. BER)while fulfilling specific constraints (on the total transmit power,maximum allowable BER). The search engine may be blind or semi-blind(i.e., some known information may be modeled into closed-form functionsand incorporated into the search to speed up running time). The searchmay be performed over both phases and amplitudes, as applicable, of eachweight coefficient. For example, in one embodiment, the phases belong toa finite range between 0 and 360 degrees where the search step may betaken between 1 and 10 degrees. The amplitudes in some embodimentsbelong to a finite range of [0, 20 dB] where the search step may betaken between 0.1 to 1 dB.

Referring next to FIGS. 5A and 5B, shown are block diagrams of twoexemplary transmitter/receiver systems 500, 550 that are capable ofadhering to the requirements of the IEEE 802.11a standard. That is, thetransmitters 508, 560 use OFDM modulation, where a stream of N_(t)consecutive quadrature amplitude modulation (QAM)-modulated datasymbols, denoted by {s₀, s₁, . . . , s_(N) _(t) ⁻¹} is modulated onto aset of N_(t) orthogonal subcarriers, see, e.g., J. Heiskala and J.Terry, OFDM Wireless LANs: A Theoretical and Practical Guide, SamsPublishing, December 2001, which is incorporated herein in its entiretyby reference.

Referring initially to FIG. 5A, shown is a block diagram of a singlechannel MIMO-OFDM system 500 using two-transmit antennas 502 _(A), 502_(B) and two-receive antennas 506 _(A), 506 _(B), in accordance with oneembodiment of the present invention. As shown, the transmitter 508 inthe present embodiment includes an encoder block 510, aserial-to-parallel converter 511, a channel state information (CSI) andmode portion 512, a weight calculation portion 514 and a signalweighting portion 516. In the present embodiment, the signal S isencoded by the encoder block 510 and then separated by theserial-to-parallel converter 511 into parallel data substreams 513. Thesignal weighting portion 516 receives and weights the parallel datasubstreams 513 with transmit signal weight values 515 received from theweight calculation portion 514.

As shown, the weight calculation portion 514 receives information aboutthe channel state and the current operating mode from the CSI and modeportion 512. Based upon the state of the channel and the transmissionmode of the signal, the weight calculation portion 514 determines thetransmit weighting values for the signal as a function of frequency.

In the exemplary embodiment, the weighted signal, comprising weightedparallel data substreams 517, is then transmitted through each of aplurality of antennas 502 _(A), 502 _(B) after being spatially weightedwith one of a corresponding plurality of antenna spatial weightingmodules 522 _(A), 522 _(B) and converted, using corresponding InverseFast Fourier Transforms 524 _(A), 524 _(B), into an OFDM signal for eachof the plurality of antennas 502 _(A), 502 _(B). It is noted that thespatial weighting 522 _(A), 522 _(B) in this embodiment is implementedat base band, and as such, the antenna (spatial) weights are availablefor each OFDM tone at both the receiver and transmitter

The signal transmitted by antennas 502 _(A), 502 _(B) then propagatesthrough the channel and is received by antenna elements 506 _(A), 506_(B) and then converted into baseband. After serial to parallelconversion 524 _(A), 524 _(E) the received baseband signals aremultiplied by receive spatial weights 526 _(A), 526 _(B), at each tone.After weighting, the signals are provided to an FFT 528 and combined.The combined received signal at the output of the FFT 528 is thendecoded 518 to generate a replica of the original signal.

Turning again to FIG. 5B, shown is a block diagram depicting a singlechannel MEMO-OFDM system 550 with two transmit 552 _(A), 552 _(B) andtwo receive 580_(A), 580 _(B) antennas in accordance with anotherembodiment of the present invention. As shown, the system in FIG. 5B isa multiple weight system where one complex RF weight 554, 578 isavailable per antenna for all the tones at the transmitter 560 andreceiver 570. In this case, the spatial weights are implemented at RF,but in alternate implementations, the RF-based weighting 554, 578 withinthe transmitter 560 and receiver 570 of FIG. 5B may be complemented bysimilar arrangements at baseband. The calculation of the spatial weightsfor both cases is described in the above-identified copending U.S.non-provisional application Ser. No. 10/801,930 for maximization of theoutput SNR, and in co-pending U.S. non-provisional Application entitledWEIGHT GENERATION METHOD FOR MULTI-ANTENNA COMMUNICATION SYSTEMSUTILIZING RF-BASED AND BASEBAND SIGNAL WEIGHTING AND COMBINING BASEDUPON MINIMUM BIT ERROR RATE, filed Jul. 13, 2004, which itself claimspriority to U.S. Provisional Application Ser. No. 60/488,845, filed Jul.21, 2003, for minimization of the output bit error rate.

In operation, a signal S is first encoded 556 and then separated by theserial-to-parallel converter 558 into parallel data substreams 559. Asignal weighting portion 562 receives and weights the parallel datasubstreams 559 with transmit signal weight values 563 received from aweight calculation portion 564.

As shown, the weight calculation portion 564 receives information aboutthe channel state and the current operating mode from the CSI and modeportion 566. Based upon the state of the channel and the transmissionmode of the signal, the weight calculation portion 564 determines thetransmit weighting values for the signal as a function of frequency.

In the exemplary embodiment, the weighted signal, comprising weightedparallel data substreams 567, is then converted, using an Inverse FastFourier Transform 568 into an OFDM signal that is up-converted to RFdomain, split and each version of the OFDM signal in the RF domain isspatially weighted 554 _(A), 554 _(B) and transmitted over acorresponding one of the transmit antennas 552 _(A), 552 _(B). It isobserved that in the embodiment of FIG. 5B, the combining weights 554are implemented in the RF domain rather than at baseband, which enablesthe number of transmit RF chains to be reduced to one.

The signal transmitted by antennas 552 _(A), 552 _(B) then propagatesthrough the channel and is received by antenna elements 580 _(A), 580_(B), and each RF signal received by the receive antennas is multipliedby corresponding receive spatial weights 578 _(A), 578 _(B) before beingcombined, converted into baseband, and converted from serial to parallelsubstreams 581, which are provided to an FFT 582 and combined. Thecombined received signal at the output of the FFT 582 is then decoded584 to generate a replica of the original signal S.

Referring next to FIG. 6, shown is a flowchart illustrating steps of thefrequency signal weighting method carried out by the transmitters ofFIGS. 4, 5A and 5B according to one embodiment of the present invention.

Initially, when the transmitter first powers up (Step 602), and thechannel state is still unknown, a set of “default” frequency signalweight values are used by the signal weighting portion 416, 516, 562.Since these transmit signal weights can only improve performance, thedefault set of weights can be chosen, for example, as if transmit signalweighting was disabled, or equivalently, all weights are set to unity.

Next, channel state information-(CSI) is acquired (Step 604). In someembodiments, operations to acquire CSI are carried out at the receiver,and the relevant information is fed back over the air, via a controlmessage, to the transmitter to the CSI and mode acquisition portion 412,512, 566 of the transmitter. In these embodiments, a training sequencecomposed of known symbols is sent from the transmitter 408, 508, 560 tothe receiver 404, 504, 570. At the receiver 404, 504, 570, the channelis estimated based on the received signal and the known sequence ofsymbols. There exists many channel estimation techniques based ontraining sequences, e.g., see J.-J. van de Beek et al., “On ChannelEstimation in OFDM Systems,” IEEE 45th Vehicular Technology Conference,vol. 2, 25-28 Jul. 1995, pp. 815-819, which is incorporated herein byreference.

Next, in some embodiments, once the channel is known, an algorithm isemployed to decide which of the possible mode candidates is best suitedto the current CSI (Step 606). The algorithm is usually referred to aslink adaptation, which ensures that the most efficient mode is alwaysused, over varying channel conditions, given a mode selection criterion(maximum data rate, minimum transmit power). Additional details on linkadaptation for frequency-selective MIMO systems are in, “AdaptiveModulation and MIMO Coding for Broadband Wireless Data Networks,” by S.Catreux et al., IEEE Communications Magazine, vol. 40, No. 6, June 2002,pp. 108-115. At this point, both channel state and mode information maybe fed back to the transmitter 408, 508, 560, and the weight calculationportion 414, 514, 564 uses this information to compute the transmitsignal weight values.

In variations of these embodiments, transmit signal weight values arealternatively calculated at the receiver and the resulting weights arefed back to the transmitter via a control message over the air. Notethat this feedback assumes that the channel varies slowly enough thatthere is sufficient correlation between the CSI used to compute theweights at the receiver and the CSI the weights are applied to at thetransmitter.

In other embodiments, all operations to establish CSI and modeacquisition are carried out at the transmitter 408, 508, 560. In certainsystems (e.g., Time Division Duplex (TDD) systems in noise-limitedenvironment) the uplink channel is the same as the downlink channel.Therefore, the transmitter may estimate the channel, compute the modeand transmit signal weight values and use those estimated parameters fortransmission over the downlink channel. In these other embodiments, thetransmitter receives a training sequence from the uplink channel,carries out channel and mode estimation and finally computes thetransmit signal weight values. This avoids the need for feedback.

After the channel state becomes available, the default weights arereplaced by more optimal frequency weights that are computed (e.g., bythe weight calculation portion 414, 514, 564) based on the current CSIand current mode (Step 608).

In the multiple carrier (OFDM) embodiments described with reference toFIGS. 5A and 5B, each tone is scaled by a transmit signal weight basedon the current CSI and current mode. The scaled data symbol at tone k isdenoted by:a _(k) s _(k)  (1.)

In some embodiments, once each tone is scaled by a transmit signalweight, then spatial weights are applied (Step 610). In the context ofOFDM systems, the scaled data symbols a_(k)s_(k) are multiplied by thetransmit spatial weights for each of the multiple transmit antennas, andthe transmitted signal out of antenna i is written as:txs _(i,k) =v _(i,k)α_(k) s _(k)  (2.)The transmit signal vector at tone k istxs _(k) =v _(k)·α_(k) s _(k)  (3.)where v _(k)=[v_(1,k), . . . , v_(n) _(T) _(,k)]^(T) is a n_(T)×1 vectorwith n_(T) as the number of transmit antenna elements.

The signal transmitted by antenna i then propagates through the channeland is received by an array of M antenna elements where it is multipliedby the receive spatial weights, denoted by u _(k)=[u_(1,k), . . . ,u_(M,k)]^(T) at each tone. After weighting, the signal is provided tothe FFT and combined. The combined received signal at the output of theFFT is written as:y _(k) =u _(k) ^(H) H _(k) ·v _(k)·α_(k) s _(k) +u _(k) ^(H) n_(k)  (4.)where H_(k) is the channel frequency response at tone k, a matrix ofsize M×n_(T) and n is complex-valued additive white gaussian noise(AWGN) with zero-mean and variance σ².

As described earlier, in certain embodiments coding and temporalequalization are not utilized, and smoothing is used to improve thereceiver performance. In OFDM-based systems, such as those describedwith reference to FIGS. 5A and 5B, the process of applying smoothingincludes scaling the tones with transmit signal weights such that thepost-processing SNR (also referred to herein as the output SNR), at thereceiver is flat across the frequency bandwidth (BW). Thepost-processing SNR corresponding to (4) is:

$\begin{matrix}{{SNR}_{k} = \frac{{{{\underset{\_}{u}}_{k}^{H}{H_{k} \cdot {\underset{\_}{v}}_{k}}}}^{2}{\alpha_{k}}^{2}{E\left\lbrack {s_{k}s_{k}^{*}} \right\rbrack}}{\sigma^{2}{{\underset{\_}{u}}_{k}}^{2}}} & (5.)\end{matrix}$When smoothing is introduced, the value of α_(k) is such that SNR_(k) isthe same at each tone. According to (5), the solution for α_(k) is:

$\begin{matrix}{\alpha_{k} = \frac{{\underset{\_}{u}}_{k}}{{{\underset{\_}{u}}_{k}^{H}{H_{k} \cdot {\underset{\_}{v}}_{k}}}}} & (6.)\end{matrix}$If the receive spatial weights are unit-norm at each tone, the solutionfor α_(k) becomes:

$\begin{matrix}{\alpha_{k} = \frac{1}{{{\underset{\_}{u}}_{k}^{H}{H_{k} \cdot {\underset{\_}{v}}_{k}}}}} & (7.)\end{matrix}$

In order to keep the total transmit power across the tones, (i.e.,N_(t)·P), constant regardless of the number of transmit antenna elementsor whether transmit frequency signal weighting is used or not, we assumethat each of the digital symbols has a power P/n_(T), i.e.,E[s _(k) s _(k) *]=P/n _(T)  (8.)

The total transmit power across the tones based on (3) and (8) is

$\begin{matrix}{{TXPW} = {{\sum\limits_{k = 1}^{N_{t}}{E\left\lbrack {\alpha_{k}^{*}s_{k}^{*}{\underset{\_}{v}}_{k}^{H}{\underset{\_}{v}}_{k}\alpha_{k}s_{k}} \right\rbrack}} = {{\sum\limits_{k = 1}^{N_{t}}{{\underset{\_}{v}}_{k}^{H}{\underset{\_}{v}}_{k}{\alpha_{k}}^{2}{E\left\lbrack {s_{k}s_{k}^{*}} \right\rbrack}}} = {{{P/n_{T}}{\sum\limits_{k = 1}^{N_{t}}{{\underset{\_}{v}}_{k}^{H}{\underset{\_}{v}}_{k}{\alpha_{k}}^{2}}}} = {N_{t}P}}}}} & (9.)\end{matrix}$

then the constraint on the frequency-scaled transmit weights isexpressed as

$\begin{matrix}{{\sum\limits_{k = 1}^{N_{t}}{{\underset{\_}{v}}_{k}^{H}{\underset{\_}{v}}_{k}{\alpha_{k}}^{2}}} = {N_{t}n_{T}}} & (10.)\end{matrix}$

Transmit smoothing as illustrated above unevenly redistributes the totaltransmit power across the bandwidth, by improving the SNR of the worsttones, while decreasing the SNR of the better tones. In the event thatthe channel goes through a deep fade in a particular tone, most of thepower will be redirected towards that particular tone, which would notbe optimal.

Thus, in one embodiment, a criterion is added to the smoothing algorithmdescribed above to limit the maximum peak transmit power applicable toone tone. In other words, the value of α_(k) is upper-bounded by athreshold. In this way, when a particular tone undergoes deep fading,that particular tone does not detrimentally draw a disproportionateamount of available power.

When coding is used, the transmit smoothing method described abovedegrades the receiver performance because of the reduced average outputSNR (as shown in the simulation results below); thus in severalembodiments, when coding is used, a transmit signal weighting techniqueother than the smoothing technique described above is utilized.

In several embodiments for example, a weighting technique referred toherein as Quantized Partial Signal Weighting (QPSW) is utilized when thesystem uses coding. With this QPSW weighting technique, the power of apercentage of tones (those corresponding to an Xth percentile of thelargest output SNR at each tone) is scaled down by an amount of A dB(where A can be a constant or a function of the output SNR) whilescaling up the transmit power of another percentage of tones (thosecorresponding the Yth percentile of smallest output SNR at each tone) byan amount of B dB (where B is a constant or a function of the outputSNR). In these embodiments, the values of X, Y, A and B are dependentupon the coding technique used. Given a code, these values may be foundwith a global search.

FIGS. 7 and 8 show the effect of smoothing on the systems of FIGS. 5Aand 5B. In particular, FIG. 7 shows the packet error rate versus receiveSNR for a signal formatted consistently with 802.11a “mode 1” (i.e. BPSKwith coding rate ½), and FIG. 8 shows the packet error rate versusreceive SNR for 802.11a “mode 10” (BPSK uncoded) signals.

Curves shown for the system described with reference to FIG. 5A arelabeled MW-BB (for multiple weights at baseband) and curves for thesystem described with reference to FIG. 5B are labeled MW-RF (formultiple weights at RF). Results for FIG. 5B are shown for both themaximum SNR criterion as well as the minimum bit error rate criterion.Selection diversity results are also shown (labeled sel).

As shown in FIG. 7, for mode 1, smoothing degrades the performancebecause mode 1 is operative using rate ½ coding. As shown, at a 10**−1packet error rate, this degradation is 0.4 dB for the system describedwith reference to FIG. 5A and 1.9 dB for the system described withreference to FIG. 5B.

In contrast, as shown in FIG. 8, for mode 10 smoothing improvesperformance because this mode is uncoded. As shown, for a 10**−1 packeterror rate, smoothing improves performance by 2.9 and 2.35 dB for thesystems described with reference to FIGS. 5A and 5B, respectively. Thus,these results validate the frequency weighting algorithm of severalembodiments, which establishes transmit signal frequency-selectiveweights as a function of the mode a particular communication system isoperating under.

FIG. 9 shows the packet error rate versus receive SNR as a result of theQPSW technique on the systems described with reference to FIGS. 5A and5B operating according to mode 6 (16QAM with coding rate ¾).

For mode 6, the QPSW technique described above was implemented withX=30, A=1.5 dB, Y═X and B as a function of the output SNR. As shown inFIG. 9, for a 10**−1 packet error rate, this transmit signal weightingapproach improves performance by 0.45 and 0.3 dB for FIGS. 5A and 5B,respectively. At the level of PER at 10**−2, the improvement from QPSWon the system of FIG. 5A is 1 dB.

In several embodiments, these weights track the variations of the CSI.For example, as soon as the CSI changes, the weights are updated aswell. For example, in packet-based systems, a training sequence isembedded at the beginning of each packet; thus the CSI is available ateach packet. In video streaming applications, packets are sentcontinuously, and since the channel is not expected to changesignificantly from packet to packet, the CSI variations can beaccurately monitored and the transmit signal weights are updatedadequately. In some of these embodiments, efficiency may be improved byonly updating the weights if the CSI has varied by more than apre-selected threshold.

In more bursty applications (e.g., Internet downloading), there may bedown periods during which no packets are transmitted. Thus, in someembodiments, if the down time is longer than the channel coherence time,the weights are re-initiated to their default values and the processeddescribed above is started again.

Those skilled in the art will readily appreciate that the presentinvention extends to single carrier systems as well. Similar to OFDMembodiments, in the single carrier embodiments, the transmit signalweighting that optimizes performance will depend on the modulation,coding, and equalization technique used. As discussed above,waterfilling is optimal when ideal coding and equalization are used, butsmoothing is optimal if no equalization or coding are used at thereceiver. Since most systems fall in between these two cases with somecoding and nonideal equalization, the optimal transmit signal weightingwill vary, and can be found, e.g., by a global search. Since themodulation and coding can vary in some systems, then the optimaltransmit signal weighting will also vary with the modulation and coding(mode) accordingly.

While the present invention has been described in detail and graphicallyin the accompanying drawings, it is not limited to such details sincemany changes and modifications recognizable to those of ordinary skillin the art may be made to the invention without departing from thespirit and the scope thereof. This includes the use of this invention inmobile, fixed, narrowband/broadband, and outdoor/indoor wirelesssystems, as well as time division duplex and frequency division duplexwireless systems.

Furthermore, the foregoing description used specific nomenclature, forpurposes of explanation, to provide a thorough understanding of theinvention. However, it is readily apparent to one skilled in the artthat the specific details are not required in order to practice theinvention. In other instances, well-known circuits and devices are shownin block diagram form in order to avoid unnecessary distraction from theunderlying invention. Thus, the foregoing descriptions of specificembodiments of the present invention are presented for purposes ofillustration and description. They are not intended to be exhaustive orto limit the invention to the precise forms disclosed, obviously manymodifications and variations are possible in view of the aboveteachings. The embodiments were chosen and described in order to bestexplain the principles of the invention and its practical applications,to thereby enable others skilled in the art to best utilize theinvention and various embodiments with various modifications as aresuited to the particular use contemplated. It is intended that thefollowing Claims and their equivalents define the scope of theinvention.

We claim:
 1. A method for communication in a user equipment, comprising:determining, by said user equipment, transmit power weights, which areto be utilized for communicating one or more wireless signals via awireless channel, as a function of frequency of a wireless signalcommunicated via said wireless channel, based on a transmission mode ofsaid wireless signal and a state of said wireless channel; determining,by said user equipment, transmit antenna spatial weights forcommunicating said one or more wireless signals via a plurality ofantennas; weighting, by said user equipment, said one or more wirelesssignals with a joint optimization of said transmit power weights and oneof said transmit antenna spatial weights; and transmitting, by said userequipment, via one of said plurality of antennas said weighted one ormore wireless signals in accordance with said transmission mode.
 2. Themethod according to claim 1, further comprising acquiring informationindicative of said state of said wireless channel when said wirelesssignal is transmitted over said wireless channel.
 3. The methodaccording to claim 1, further comprising acquiring informationindicative of said transmission mode of said wireless signal when saidwireless signal is transmitted over said wireless channel.
 4. The methodaccording to claim 1, wherein said determined transmit power weights ofany frequency of said weighted one or more signals are upper-bounded bya threshold.
 5. The method according to claim 1, wherein said jointoptimization takes place using a global search.
 6. The method accordingto claim 1, wherein said weighting of said one or more signals takesplace during one or both of baseband signal processing and/or RF signalprocessing.
 7. The method according to claim 1, further comprisingdetermining said transmit power weights based at least in part onreducing power of at least a portion of frequencies of said one or morewireless signals that correspond to frequencies of said wireless signalwith a signal-to-noise ratio (SNR) above a threshold.
 8. The methodaccording to claim 1, further comprising determining said transmit powerweights based at least in part on increasing power of at least a portionof frequencies of said one or more wireless signals that correspond tofrequencies of said wireless signal with a signal-to-noise ratio (SNR)below a threshold.
 9. The method according to claim 1, wherein said oneor more wireless signals comprise at least one of a single carriersignal, a GSM signal, a multi-carrier signal, a code division multipleaccess (CDMA) signal, an ultra-wideband signal, and/or an orthogonalfrequency division multiplexed (OFDM) signal.
 10. The method accordingto claim 1, wherein said transmit power weights comprise at least onescalar value.
 11. The method according to claim 1, further comprisingjointly optimizing at least one performance measure based on saidweighting of said one or more wireless signals.
 12. The method accordingto claim 11, wherein said performance measure comprises one or both ofan output signal-to-noise ratio (SNR) and/or an output packet error rate(PER).
 13. A user equipment for communication, comprising: one or moreprocessors and/or circuits configured to: determine transmit powerweights, which are to be utilized for communicating one or more wirelesssignals via a wireless channel, as a function of frequency of a wirelesssignal communicated via said wireless channel, based on a transmissionmode of said wireless signal and a state of said wireless channel;determine transmit antenna spatial weights for communicating said one ormore wireless signals via a plurality of antennas; weight said one ormore wireless signals with a joint optimization of said transmit powerweights and one of said transmit antenna spatial weights; and transmitvia one of said plurality of antennas said weighted one or more wirelesssignals in accordance with said transmission mode.
 14. The userequipment according to claim 13, wherein said one or more processorsand/or circuits are further configured to acquire information indicativeof said state of said wireless communication channel when said wirelesssignal is transmitted over said wireless channel.
 15. The user equipmentaccording to claim 13, wherein said one or more processors and/orcircuits are further configured to acquire information indicative ofsaid transmission mode of said wireless signal when said wireless signalis transmitted over said wireless channel.
 16. The user equipmentaccording to claim 13, wherein said determined transmit power weights ofany frequency of said weighted one or more signals are upper-bounded bya threshold.
 17. The user equipment according to claim 13, wherein saidjoint optimization takes place using a global search.
 18. The userequipment according to claim 13, wherein said weighting of said one ormore signals takes place during one or both of baseband signalprocessing and/or RF signal processing.
 19. The user equipment accordingto claim 13, wherein said one or more processors and/or circuits arefurther configured to determine said transmit power weights based atleast in part on reducing power of at least a portion of frequencies ofsaid one or more wireless signals that correspond to frequencies of saidwireless signal with a signal-to-noise ratio (SNR) above a threshold.20. The user equipment according to claim 13, wherein said one or moreprocessors and/or circuits are further configured to determine saidtransmit power weights based at least in part on increasing power of atleast a portion of frequencies of said one or more wireless signals thatcorrespond to frequencies of said wireless signal with a signal-to-noiseratio (SNR) below a threshold.
 21. The user equipment according to claim13, wherein said one or more wireless signals comprise at least one of asingle carrier signal, a GSM signal, a multi-carrier signal, a codedivision multiple access (CDMA) signal, an ultra wideband signal, and/oran orthogonal frequency division multiplexed (OFDM) signal.
 22. The userequipment according to claim 13, wherein said transmit power weightscomprise at least one scalar value.
 23. The user equipment according toclaim 13, wherein said one or more processors and/or circuits arefurther configured to jointly optimize at least one performance measurebased on said weighting of said one or more wireless signals.
 24. Theuser equipment according to claim 23, wherein said performance measurecomprises one or both of an output signal-to-noise ratio (SNR) and/or anoutput packet error rate (PER).